The invention relates to a method for making a multistage single-sideband shifter. This is done by cascading switchable phase shifters, because the suppression of the undesired sideband increases with an increasing number of cascaded phase switches. If the number of phase switches is N, then the effective number of stages of the single-sideband shifter, if it differs from the number of phase switches, is N.sub.eff.
Single-sideband shifters of this kind are used for instance to make homodyne network analyzers. In that case, as shown in FIG. 1, an oscillator signal 1 of frequency f.sub.1 is split via a signal divider 2 into a measurement signal branch 3 and a reference signal branch 4. In measurement signal branch 3 the N switchable phase shifters, here for instance three phase shifters 5, 6 and 7, and the measurement target, or device being tested, 8 are connected in series to produce an output signal at frequency f.sub.2. The signals of reference branch 4 and measurement branch 3 are superimposed in a mixer 9, producing an intermediate frequency signal of frequency f.sub.O =.vertline.f.sub.2 -f.sub.1 .vertline., from which possible secondary waves, a term which includes for instance harmonic waves or subharmonics, are eliminated in a bandpass filter 10.
Each device 5, 6 and 7 can be a conventional phase shifter having a bypass switch connected between its input and output so that when the switch is closed, a conductive path is established between the input and output and no or a small phase shift takes place. Each bypass switch can be connected to a control circuit which controls its switching state.
The device under test, 8, may be any active or passive two-port device, e.g., an electrical filter, an amplifier, a switching network, an antenna arrangement, etc.
If the single-sideband shifter is operated in accordance with the invention, which means that the frequency of the frequency-shifted useful signal is f.sub.2 =f.sub.1 .+-.f.sub.O, and that the associated image signal representing interference is quite low, then the bandpass-filtered intermediate frequency signal, the magnitude and phase of which can be evaluated by conventional methods, is proportional to the signal of measurement branch 3 and hence is also a standard for the complex transfer function H of measurement target 8. To this end, as is well known, the mixer must be operated quasi-linearly, which means that mixer 9 must function like an ideal multiplier and/or the signal in reference branch 4 must have a large amplitude compared with the signal in measurement branch 3.
If the phase switches are initially switched very slowly, so that quasi-stationary operation prevails (f.sub.2 =f.sub.1), then the mixer, at its low-frequency output, furnishes a direct voltage that is a standard for a real portion of the complex transfer function H.sub.O =K.sub.O H of the entire signal branch. K.sub.O contains not only the basic damping and the basic phase but also the complex conversion factor of the mixer.
Also appearing at the mixer output is a direct voltage, the so-called mixer offset voltage, which is assumed to be eliminated or rendered ineffective in a suitable manner. The way this happens will be described in detail hereinafter. With these preconditions, the following equation is applicable for the output voltage of the mixer: EQU U.sub.ml =U.sub.1 =ReH.sub.O =1/2(H.sub.O +H.sub.O.sup.*),
where H.sub.O.sup.* designates the complex variable conjugated for H.sub.O.
First, as an example, an apparatus including two phase switches will be considered, in which the phase switches are decoupled, so that their switchover can be completely described by a complex factor k. The entire transfer function H.sub.O is varied from H.sub.O to H.sub.O k.sub.1 by switching on the phase switch I; similarly, for switching on the phase switch II, H.sub.O .fwdarw.H.sub.O k.sub.2. If both phase switches are turned on simultaneously, the result is H.sub.O .fwdarw.H.sub.O k.sub.1 k.sub.2, so that the associated mixer output voltages become as follows: ##EQU1##
These voltages are linearly superimposed in a suitable manner, so that with the definition: EQU .rho..sub.1 =1, .rho..sub.2 =p.sub.1, .rho..sub.3 =p.sub.2 and .rho..sub.4 =p.sub.1 p.sub.2
for the sum voltage U.sub.1, the result is ##EQU2## which if the weighting factors p.sub.1 and p.sub.2 are suitably selected, namely in accordance with equation 6, ##EQU3## is now proportional only to the desired transfer function H, but no longer to H*. This is because, on the condition of equation 6, it is true that: ##EQU4## in which the proportionality constant ##EQU5## is determined by a calibration measurement without a measurement target.
Generally, that is, with N phase switches, 2.sup.N different switching states are possible, so that the voltages U.sub.i, where . . . i=1 . . . 2.sup.N, are measured, which are summed up to make the sum voltage U.sub.1 in accordance with the following: ##EQU6## which on the following conditions ##EQU7## is simplified to ##EQU8##
However, if the weighting factors are not selected precisely in accordance with equation 10, then the sum voltage U.sub.1 includes interference H.sub.O.sup.* - although less so, the more factors the products in equation 9 have, because the interference term then diminishes increasingly exponentially, since even with deviations from the condition of equation 10, it is still always true that (1+p.sub.i k.sub.i.sup.*)&lt;&lt;1. Since even in the absence of decoupling the phase switch has the effect of a deviation from the exact weighting factors, with an increasing number of stages it is increasingly possible to omit the decoupling.
The evaluation of the mixer output voltages according to equation 8 can be performed in that the U.sub.i is measured sequentially and summed up in a calculatedly weighted manner. For high-speed applications, however, this kind of procedure is too time-consuming; this is primarily due both to the transient events in the filter and to the finite conversion rate of the digital/analog converters.